Partial response demodulating method and apparatus using the same

ABSTRACT

In a data demodulating method, predetermined input data is demodulated based upon a response characteristic of the partial response class 4; the demodulated input data is discrete-filtered to thereby produce filtering data; and the filtering data is maximum likelihood-decoded to thereby producing asymmetrical response data. Further, a magnetic recording/reproducing apparatus is arranged by using this data demodulating method.

CROSS-REFERENCE TO RELATED APPLICATION

[0001] The present application relates to subject matter described inU.S. application Ser. No. 08/975,670, filed Nov. 28, 1997 entitled“INFORMATION RECORDING/REPRODUCING METHOD AND APPARATUS USING EPRMLCONNECTION PROCESSING SYSTEM” now U.S. Pat. No. 6,069,856, thedisclosure of which is hereby incorporated by reference.

BACKGROUND OF THE INVENTION

[0002] The present invention is generally related to a signal processingsystem for either a magnetic disk apparatus or an optical diskapparatus. More specifically, the present invention is directed to ahigh-efficiency demodulating method of a high-order partial responsesystem such as an EEPRML (Extended Extended Partial Response MaximumLikelihood) signal processing system and an EEEPRML (Extended EEPRML)signal processing system.

[0003] In magnetic disk apparatuses, the partial response maximumlikelihood (will be abbreviated as a “PRML” hereinafter) signalprocessing system combining the partial response class 4 (PR4) and themaximum-likelihood decoding system are practically available as ahigh-efficiency signal processing system. A high-efficiency signalprocessing system implies a system capable of realizing a desirable dataerror rate at a low SIN (signal-to-noise) ratio. Very recently, suchhigh order partial response systems have been practically utilized assignal processing systems capable of reproducing signals at SIN ratioslower than that of the PRML system, for instance, the EPRML system bycombining the EPR4 (Extended PR4) with the maximum (most) likelihooddecoding system.

[0004]FIG. 1 shows a structural example of the construction of a generalmagnetic disk apparatus using a PRML signal processing system. originaldata is supplied to an error correcting encoder 7 through an interfacecircuit 8 so that the original data is added with redundant datanecessary for error correction. Next, the original data added withredundant data is subjected by a data modulator 6 to modulationnecessary for the PRML system and is recorded on a magnetic disk 3 by amagnetic head 4 through a recording/reproducing amplifier 5. A signalreproduced from the magnetic disk 3 is passed through therecording/reproducing amplifier 5 and then PRML-processed by a datademodulator 1. The demodulated data is error-corrected by an errorcorrecting decoder 2 and is thereafter converted through the interfacecircuit 8 into the original data. The operation and the arrangement ofthis data modulator 6 and data demodulator 1 will now be explained inmore in detail with reference to FIG. 2 indicating a relationshipbetween a magnetic recording/reproducing system and a partial responsesystem. A first description will now be made of process operationsexecuted on the data recording side. The data outputted from the errorcorrecting encoder 7 is penetrated through a precoder 9 constructed of adelay element and modulo (Mod. 2), and then is recorded via a recordingamplifier 5 on a recording medium. This precoder 9 is employed so as toprevent erroneous propagation of data which is caused during thedemodulating operation.

[0005] Next, a description will now be made of a processing operation onthe reproducing side. The magnetization on the recording medium isreproduced as a waveform having a differential characteristic by therecording/reproducing head. PR4 may regard this differentialcharacteristic as a differential system of (1-D). In this case, symbol“D” implies a 1-bit delay calculator. The reproduced waveform issupplied to the equalizer 10 so as to be equalized in such a manner thata response of the waveform becomes (1+D). As a result, a total transfercharacteristic in the output of the equalizer becomes (1−D²).Thereafter, a data discrimination of the data is carried out in themaximum decoder 11. FIG. 3 represents a response of a regenerativeisolated waveform (note that a step response will be simply referred toas an “isolated waveform” hereinafter) in the case that a step waveformis magnetically recorded. PR4 implies that the isolated waveform isregarded as a waveform enlarged to 2 time slots, as indicated in FIG.3A. This waveform owns such a characteristic having (1+D). Also, asindicated in FIG. 3B, EPR4 implies that the isolated waveform isregarded as a waveform enlarged to 3 time slots. This waveform owns sucha characteristic having (1+D)². Furthermore, as indicated in FIG. 3C,EEPR4 implies that the isolated waveform is regarded as a waveformenlarged to 4 time slots. This waveform owns such a characteristichaving (1+D)³.

[0006] While considering the EEPR4 system as an example, the high orderpartial response system will now be summarized.

[0007] A total transfer characteristic of EEPR4 constitutes (1−D)·(1+D)³as a product of a transfer characteristic of an isolated waveform andanother transfer characteristic of a magnetic recording system. Animpulse response of the EEPR4 system determined by this product isrepresented in FIG. 4. As apparent from a waveform “a” shown in FIG. 4,an isolated waveform of EEPR4 owns amplitude characteristics (normalizedratio) of 1, 3, 3, 1 every bit period. As a consequence, as indicated ina waveform “b” of FIG. 4, a response of an isolated pulse is obtained bysuperimposing the isolated waveforms inverted along the upper/lowerdirections with each other by shifting a 1-bit time period. In otherwords, the response of the isolated pulse becomes 1, 2, 0, −2, −1. InFIG. 5, there is shown a trellis diagram of EEPRML obtained by combininga maximum likelihood decoder with EEPR4. As is well known in this field,the operation of the EEPRML system may be explained based upon thetrellis diagram. In the drawing, symbol “ak” indicates an input signalto EEPRML at a time instant “k”. In this case, reference numeral 12indicates a state, and reference numeral 13 shows a state transition. Anupper stage of a label (ak/yk) and a lower stage thereof indicate aninput signal value and an output signal value, respectively. The statesof the respective signal processing systems are determined by the pastinput signal series. In EEPRML, a level of a reproduction signal at thepresent time instant is influenced by signals over the past 4 timeslots. Assuming now that a state at a time instant “k” is equal to “sk”,it is given as

sk=((ak−4, ak−3, ak−2, ak−1)1ak(1,0)),

[0008] and a total number of states becomes 16. At a time instant “k−1”,state transitions originated from a plurality of states are collected toa specific state at the time instant “k”. With respect to these statetransitions, a squared value of a difference between an output signaland an input signal, which are indicated at a low stage of each label,will be referred to as a “branch metric”. Also, an accumulated value ofbranch metrics until the present time instant with respect to each ofthe states will be referred to as a “path metric”. Among the statetransitions collected to a specifie-state at the time instant “k”, onlysuch state transitions that a summation becomes a minimum value and thissummation is made from path metrics until a time instant “k−1” andbranch metrics corresponding to the respective state transitions areselected as a state transition (path) capable of satisfying a maximumlikelihood condition (most certain). This stage may be subdivided intothe below-mentioned steps. In other words, the path metrics are added tothe branch metrics (Add). Next, these added values are compared witheach other every state, and such a state transition which becomes aminimum value is selected (Select). A series of these operations will beabbreviated as an “ACS”. The maximum (most) likelihood decoding methodis such well-known method that this ACS operation is repeatedlyperformed at each time instant and under each state, and then when thepath metrics are finally converged onto one path on the trellis diagram,the data is determined.

[0009] The performance of EEPRML is determined by a minimum freedistance (Dfree). In this case, “Dfree” implies a minimum difference ofpath metrics among various sorts of combinations from a specific node toanother specific node on the trellis diagram shown in FIG. 5. It isknown that “Dfree” of EEPRML is equal to 6. Furthermore, distancesbetween signals subsequent to “Dfree” become 8 and 10. These distancesbetween signals of EEPRML are determined by a data pattern entered intothe maximum likelihood decoder. In particular, a distancebetween-signals is defined by a continuous time at which a pattern ischanged from 0 to 1, or from 1 to 0. As will be discussed later,assuming now that an inverting position contained in a pattern isexpressed by, for example, “p”, in such a case that 2 sorts of patternsare set under 1-bit shifted condition, and these patterns own 3-timecontinuous inverting positions such as “p+p”, a distance between thesepatterns may give “Dfree”. To further improve the performance of theseEPRML system and EEPRML system, very recently, Maximum Transition RunCode (will be abbreviated as an “MTR code” hereinafter) has beenproposed.

[0010] For instance, the conventional MTR code is described in, forexample, “Maximum Transition Run Codes for Data Storage Systems”, IEEETransactions on Magnetics, volume 32, No. 5, September 1996, pages 3992to 3994. The above-described MTR code owns a function to restrict thatinverting of a pattern occurs more than 3 times. When this MTR code isused, a limitation can be made to the pattern inversion for more than 10distances between signals of EEPRML. As a consequence, an S/N ratio of asignal can be equivalently improved. However, in the MTR code, the coderate becomes 4/5 and the like. This code rate value is low, as comparedwith the normally used 16/17 GCR (Group Coded Recording) and 8/9 GCR. Asa result, a code rate loss becomes large, and a total coding gain cannotbe always satisfied. Concretely speaking, a gain becomes approximately2.2 dB, since the distance between signals is improved-from 6 to 10. onthe other hand, a code rate loss becomes larger than approximately 1 dBunder normalized line density=3, for instance, (normalized linedensity=a half bandwidth of a reproduced waveform is normalized by awidth of a recording pulse), and a total coding gain becomes at maximumapproximately 1 dB, depending upon a recording density of a magneticdisk.

SUMMARY OF THE INVENTION

[0011] An object of the present invention is to provide a generally-usedmethod for expanding the distance between signals of a high orderpartial response system, especially, the EEPRML system and the EEEPRMLsystem irrespective of a code under use. In other words, an object ofthe present invention is to provide a method for equivalently expandinga distance between signals without newly producing a code rate loss,since the 16/17 GCR, or

[0012] the 8/9 GCR used in the PRML signal process operation for amagnetic disk apparatus can be directly applied.

[0013] In accordance with the present invention, in the high orderpartial response system, especially the EEPRML system and the EEEPRMLsystem, a response of an isolated pulse waveform is changed from theoriginal response of EEPRML, or EEEPRML, so that the distance betweensignals can be expanded. In the high order partial response system, aresponse of an isolated pulse is selected to be an symmetrical waveform.For instance, as previously described, in the EEPRML system, theresponse of the isolated pulse waveform becomes 1, 2, 0, −2, −1.

[0014] In accordance with the present invention, since the asymmetricalcharacteristic owned by the response of the isolated pulse waveform insuch a high order partial response system is relaxed, the distancebetween signals is firstly extended. This distance between signalsdetermines the S/N ratio when the signal is discriminated. it is impliedthat the longer this distance between signals becomes, the larger theamplitude of the signal equivalently becomes. Secondly, the noise poweris reduced. The noise of the partial response owns correlativerelationships with each other over plural time instants. The performanceof the maximum likelihood decoder is deteriorated by the adverseinfluence caused by this correlative relationship of the noise. As aresult, the noise can be essentially reduced by suppressing thecorrelative characteristic of the noise. In other words, an S/N ratio ofa high order partial response signal is defined by the followingformula:

S/N=distance between signals/(noise power×noise correlativecoefficient)  (1).

[0015] While considering the EEPRML system, the present invention willnow be explained with a concrete example. A code is equal to a binarynumber of {1, 0}. Now, in order to defame a dimension of a code error, avalue of 1 corresponds to such an error case that 1 erroneously becomes0; a value of −1 corresponds to such an error case that 0 erroneouslybecomes 1; and a value of 0 corresponds to such an error case that noerror occurs. The error patterns of the EEPRML system are classified inaccordance with this definition:

[0016] (A). In a case that the distance between signals=6 (1, −1, 1).

[0017] (B). In a case that the distance between signals=8:

[0018] 1) (1, −1,1,0,0,1,−1,1)

[0019] 2) (1,−1,1,−1,1).

[0020] (C). In a case that the distance between signals=10: (0, 1, 0)etc.

[0021] An actual code error pattern of (A) is such a case that (a, b, 1,0, 1, c, d) erroneously becomes (a, b 0, 1, 0, c, d), or vice versa.

[0022] An actual code error pattern of (B) 1) is such a case that (1, 0,1, a, b, 1, 0, 1) erroneously becomes (0, 1, 0, a, b, 0, 1, 0), or viceversa.

[0023] An actual code error pattern of (B) 2) is such a case that (1, 0,1, 0, 1, 0, 1) erroneously becomes (0, 1, 0, 11 0, 1), or vice versa. Inthis case, symbols “a”, “b”, “c”, and “d” are arbitrary.

[0024] (C) is a 1-bit isolated pulse error.

[0025] As previously described, with respect to the patterns commonlyapplied to (A) and (B), the signals are inverted at least 3 times. As aconsequence, in the data pattern, either “ab1010cd” or “ab0101cd”, andalso these data patterns are continued. FIG. 6 represents that the errorof the distance between signals=6 is plotted on a trellis diagram. Twosorts of data streams

shown in FIG. 6 own such values of “010abcde” and “101abcde”. That is,only 3 bits thereof are different from each other. FIG. 7 indicateswaveforms corresponding to these data streams. As apparent from thisdrawing, a distance between signals of the two sorts of patterns is 6.Similarly, FIG. 8 represents a waveform corresponding to theabove-described (B) 1).

[0026] On the other hand, EEPRML has a transfer characteristic of(1−D)·(1+D)3. As a result, as illustrated in the waveform “b” of FIG. 4,an impulse response is determined by 1, 2, 0, −2, −1. Therefore, in thecase that a 1-bit error happens to occur, a distance between signals forthe erroneous pattern and the pattern originally having no error becomes10, namely equal to a squared summation of the respective values of thisimpulse response. This distance between signals is equal to energyitself owned by this impulse signal. As a consequence, the reason whythere is such a pattern of the distance between signals=6, or thedistance between signals=8, as indicated in FIG. 7 and FIG. 8 is givenas follows. That is, a combination of these patterns is to cancel theenergy owned by the original impulse signal. In other words, the EEPRMLsystem involves such a pattern that an erroneous propagation readilyoccurs. A further consideration will now be made of the cause why such adistance between signals is decreased with reference to a waveformdiagram shown in FIG. 9A and an impulse response indicated in FIG. 9B.This diagram shows a pattern whose distance between signals becomes 6 inEEPRML indicated in FIG. 7. In this pattern, as shown in FIG. 9A,inverting of the signal is continued 3 times, namely P1, P2, P3. As aresult, as indicated in FIG. 9B(a), isolated waveforms having responses1, 3, 3, 1 are alternately repeated in such a manner ofpositive-negative-positive. As a consequence, such a response of 1, 2,1, 1, 2, 1 is obtained, and energy of-this signal becomes a squaredsummation of the respective values, namely(1)²+(2)²+(1)²+(1)²+(2)²+(1)²=12. On the other hand, signal energy ofeach of isolated waveform single body becomes (1)²+(3)²+(3)²+(1)²=20.Accordingly, in such a pattern that inverting of signals is continued 3times, namely P1, P2, P3, a total signal energy of the 3 isolatedwaveforms, i.e., 60 is reduced to 12. For example, as shown in FIG.9B(b), when the response of the isolated waveform having the amplitudeof 1 located at the right end is eliminated from the responses of theisolated waveforms, the distance between signals may be increased up to15. This implies that since the responses 1, 3, 3, 1 of the isolatedwaveforms in EEPRML are excessively extended to a plurality of bits, theoriginal energy of the signal is canceled in the specific patterns shownin-the above-described (A) and (B). This reason may cause the distancebetween signals to be essentially reduced. As a result, the erroneouspropagation is induced.

[0027] Based upon this consideration, the essential aspect in order toenlarge the distance between signals is to establish a measure how toconcentrate energy without losing the energy (electric power) of theisolated waveform. In general, as shown in FIG. 10, a means forconcentrating energy of a signal an isolated waveform is filtered by anall-pass filter 14 to satisfy a minimum, phase transition condition,which could be cleared based on the communication theory. In this case,a minimum phase transition condition implies that a zero point and apole of a transfer function of a signal given by a rational function arepresent within the same unit circumference.

[0028] Since a phase filter is set so as to satisfy this condition,energy of a signal can be concentrated to a front half portion of animpulse response while the energy of this signal is reserved. In amagnetic recording operation, it is well known that an isolated waveformcan be approximated by way of the Lorentz waveform. When this is givenby L(t), it is expressed by the below-mentioned formula (2):

L(t)=1.O/(l+(2t/TW)2)  (2)

[0029] where symbol “TW” gives a half bandwidth.

[0030] As apparent from this formula (2), L(t) is a symmetrical waveformwith respect to right/left directions. In this case, a ratio (TW/T) of ahalf bandwidth to a time width T of a pulse to be recorded is defined asa normalized line density. When the value of this ratio TW/T isincreased, the waveform may be recorded in the high density. Normally,in the magnetic recording operation, such a Lorentz waveform whosenormalized line density is selected to be approximately 2.5 is used. Itis now assumed that waveforms produced by filtering a Lorentz waveformhaving a normalized line density of 2.5 and another Lorentz waveformhaving a normalized line density of 3.0 by a minimum phase transitionfilter is recognized as L min(t), FIG. 11 indicates L min(t). Asapparent from FIG. 11, the waveforms are symmetrical with each otheralong the right/left directions. Also, it can be understood that theenergy is concentrated to the front half portion of the isolatedwaveform. However, generally speaking, it is very difficult to extract aclock signal (timing signal) required to discriminate a signal from anasymmetrical waveform. As one of these reasons, a jitter

[0031] component (temporal fluctuation) depending upon a pattern isincreased due to a phase distortion. As another reason, since the signalamplitude has multi-values, the clock signal (timing signal) extractingcircuit becomes complex. This practical reason makes it difficult. As aconsequence, according to the present invention, in order to solve—thiscontradictory condition, the polynomial PR(D) of the high order partialresponse is factorized in accordance with the following formula (3):

PR(D)=(1−D2)·(C ₀ +C,D+ . . . +C _(n) D _(n))  (3)

[0032] The above-described asymmetric characteristic is given to thewaveform by such a manner that the timing extraction is carried outunder condition of a front term in a right hand, and thereafter anasymmetrical response given by a rear term in the right hand is given bythe discrete time filter. At this time, a selection is made of suchasymmetrical coefficients C₀, C₁, . . . , C_(n) that the S/N ratio givenby the formula (1) becomes maximum.

[0033] Next, a description will now be made of an actual method forcalculating the asymmetrical coefficients. In a first case of 16 statesin EEPRML, the above-explained asymmetrical coefficients are given as(C₀=1, C₁=2, C₂=1). In other words, the value of C₀ and the value Of C₂constitute symmetrical coefficients, while setting C₁ as a center. Tothe contrary, in order to calculate the asymmetrical coefficient, firstof all, a rear term of a right hand of the formula (3) is set as mornicpolynomial of C₀=1. Assuming now that the coefficients C₁ and C₂

[0034] are regarded as 2 variable functions of a real number, an optimumcoefficient is calculated in accordance with the evaluation basis of theformula (1). Therefore, an integer coefficient most approximated to thisreal number is calculated. It should understood that since the methodsfor calculating the distance between signals, the noise power, and thenoise correlative coefficient indicated in the formula (1) are describedin detail in the publication “Maximum Likelihood Sequence Estimation ofDigital Sequences in the Presence of Intersymbol Interference”, IEEETransactions on Information Theory, vol. IT-18, No. 3, May 1972, pages363 to 378, descriptions thereof are omitted. A table 1 represents atypical characteristic of such a partial response that a state numberthereof is 16. A distance of an isolated pulse indicated in this table 1is directly equal to electric power owned by the isolated pulse itself.A minimum distance corresponds to such minimum distance among distanceson a trellis diagram of partial response signal having givencoefficients. As a consequence, a distance of the minimumdistance/isolated pulse may constitute an index for there is aparticular improvement in the characteristic. It should also be notedthat the characteristics of the table 1 and the table 2 correspond tosuch a case that the normalized line density is 2.5. Furthermore,according to the present invention, not only the S/N ratio can beimproved, but also the length of the code error can be improved, ascompared with the long continued errors caused by the conventionalEEPRML and EEEPRML systems. That is, giving the utilization efficiencyof the energy of the partial response for giving this distance. Thepartial response system having the coefficient according to the presentinvention may have the advantages as to this point, as compared withthat of the normal EEPRML. As a result, it can be seen that the S/N canbe effectively improved with respect to the EEPRML having thesymmetrical coefficient. A table 2 represents a typical characteristicof such a partial response that a state number thereof is 32. Also, inthis case, according to the present invention, the major error bitlength is the 1-bit error bit length, or the 3-bit error bit length. Asa consequence, the present invention has such a feature that the errorcorrection can be effectively performed by combining with the errorcorrection code having the code error correcting capability with respectto at least the 1-bit continuous error, and the 3-bit continuous error.

BRIEF DESCRIPTION OF THE DRAWINGS

[0035] For a more better understanding of the present invention,reference is made of a detailed description to be read in conjunctionwith companying drawings, in which:

[0036]FIG. 1 is a structural diagram for indicating the conventionaldata demodulating circuit;

[0037]FIG. 2 is a schematic diagram for showing the relationship betweenthe PRML demodulating system and the magnetic recording/reproducingsystem;

[0038]FIG. 3A, FIG. 3B, and FIG. 3C are graphic representations forshowing the conventional isolated waveform response of the partialresponse;

[0039]FIG. 4 is a graphic representation for indicating the conventionalisolated wavefortn of EEPR4 and the conventional isolated pulseresponse;

[0040]FIG. 5 represents the conventional trellis diagram of EEPRML;

[0041]FIG. 6 shows such a diagram that the conventional pattern givingthe distance between signals “6” of EEPRML is indicated on the trellisdiagram;

[0042]FIG. 7 indicates an example of waveforms giving a distance betweensignals 6 of EEPRML;

[0043]FIG. 8 indicates an example of waveforms giving a distance betweensignals 8 of EEPRML;

[0044]FIG. 9A and FIG. 9B are diagrams for indicating the reason why theconventional distance between signals of EEPRML is reduced to 6;

[0045]FIG. 10 schematically represents a basic idea for concentratingenergy of isolated waveform responses of a partial response according toan embodiment of the present invention;

[0046]FIG. 11 is a graphic representation for representing an example ofa minimum phase transition waveform according to an embodiment of thepresent invention;

[0047]FIG. 12 is a schematic block diagram for showing a circuitarrangement according to an embodiment of the present invention;

[0048]FIG. 13A and FIG. 13B are schematic block diagrams for indicatinga circuit arrangement of a discrate time filter according to anembodiment of the present invention;

[0049]FIG. 14 is a trellis diagram having coefficients of the embodimentof the present invention;

[0050]FIG. 15 schematically represents a 16-state maximum likelihooddecoder as an example of the embodiment of the present invention; and

[0051]FIG. 16 schematically indicates a data demodulating method of amagnetic disk apparatus according to an embodiment of the presentinvention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0052] In FIG. 12, there is shown a structural example of an actualcircuit arrangement according to the present invention. First, an outputof a magnetic head is supplied via a preamplifier to an AGC (automaticgain control circuit) and LPF (low-pass filter) 15. After this magnetichead output is controlled by the AGC/LPF 15 in such a manner that anamplitude of a signal becomes a constant, noise components other than adesirable frequency range are removed by this AGC/LPF 15. This LPFoutput signal is discrete-quantized by an ADC 16, and then thediscrete-quantized signal is inputted into an equalizer 10. Aspreviously explained, in the equalizer 10, the reproduction signalderived from the magnetic head is equalized in such a manner that thisreproduction signal has a partial response characteristic of (1−D²). Aclock signal required to operate the ADC 16 is produced from the outputsignal of this equalizer 10 by a PLL circuit 20. At the same time, acontrol signal of the AGC/LPF 15 is also obtained from an AGC controlcircuit 21. Next, an output signal of the equalizer 10 is applied to adiscrete time filter 18 so as to produce such a filter output signalhaving a response characteristic of (1−D²) (C₀+C₁D+ . . . +C_(nD) ^(n)).Then, this filter output signal is supplied to a maximum likelihooddecoder 19 so as to discriminate data. This discriminate data isdemodulated by a 16/17 (or 8/9) ENDEC 23 to obtain original user datafrom an output of this 16/17 ENDEC 23. It should be understood thatsince the output si-gnal of the equalizer 10 is supplied to a maximumlikelihood decoder 22 of PR4, the normal PRML demodulation data isobtained. Next, an arrangement of the discrete time filter is indicated.

[0053]FIG. 13A is a structural example of a discrete time filter havingsuch a coefficient of (C₀=3, C₁=2, C₂=1). The output of the equalizer 10is added to an input terminal 30 of the discrete time filter. An outputobtained by processing this signal by a 3-time coefficient multiplier31, another output obtained by delaying this signal by 1 bit in a delaycircuit 36 to process the 1-bit delayed signal by a 2-time coefficientmultiplier 32, and another output obtained by delaying this signal by 2bits to process the 2-bit delayed signal by a 1-time coefficientmultiplier 33 are added by an adder 34, so that a desirable filterCoefficient is obtained at an output terminal 35. As a result, a pulseresponse is given as 3, 2, −2, −2, −1

[0054] based on a formula (3). Apparently, as to other coefficients of16 states, a discrete time filter may be similarly constructed byemploying coefficients represented in a table 1.

[0055]FIG. 13B is a structural example of a discrate time filter havingsuch a coefficient of (C₀=2, C₁=5, C₂=3, C₃=2). The output of theequalizer 10 is added to an input terminal 60 of the discrete timefilter. An output obtained by processing this signal by a 2-timecoefficient multiplier 51, another output obtained by delaying thissignal by 1 bit in a delay circuit 56 to process the 1-bit delayedsignal by a 5-time coefficient multiplier 52, another output obtained bydelaying this signal by 2-bits to process the 2-bit delayed signal by a3-time coefficient multiplier 53, and another output obtained bydelaying this signal by 3 bits to process the 3-bit delayed signal by a2-time coefficient multiplier 54 are added by an adder 55, so that adesirable filter coefficient is obtained at an output terminal 56. As aresult,. a pulse response is given as 2, 5, 1, −3, −3, −2 based on aformula (3). Apparently, as to other coefficients of 32 states, adiscrete time filter may be similarly constructed by employingcoefficients represented in a table 2.

[0056] Next, there is shown a method for constituting a trellis diagramaccording to the present invention. As to a value a_(k) of an input bitto the maximum likelihood decoder, the respective states SK, and theOutput Y_(k), the 5 below-mentioned relationship defined by thefollowing formula (4) is established:

SK=a _(k)−5,^(a) k−4,^(a) k−3,^(a) k−2,^(a) k−1

Y _(k) =C0^(a) k+C ₁ ^(a) k−1+(C ₂ −C ₀)^(a) k−2+(C ₃ −C ₁) ^(a)k−3^(−c)2^(a) k−4^(−c)3^(a) k−5  (4)

[0057] The maximum likelihood decoder has 16 states in the case of C₃=0,and also has 32 states in the case that the value of C₃ is not equal tozero. In FIG. 14, there is shown a structural example of a trellisdiagram of a 16-state maximum likelihood decoder having such a value of(C₀=3, C₁,=2, C₂=1). In this case, a partial response having such acoefficient is referred to as an “MEEPRML”. FIG. 15 schematicallyrepresents one embodiment mode of the 16-state maximum likelihooddecoder Of FIG. 14. This processing circuit is arranged by a branchmetric generating unit 40, an ACS circuit 41, and a path memory 42. Thisprocess circuit is arranged based upon the MEEPRML trellis diagramindicated in FIG. 14. The branch metric generating unit 40 is to apply abranch metric of a state transition generated from each of the states inthe 215 MEEPRML trellis diagram.

[0058] The ACS circuit 41 executes an adding process, a comparingprocess, and a selecting process between the path metric values and thebranch metric values of the 16 states, so that a path metric value withrespect to a most likelihood path is generated. The path memory 42produces decoded data based upon the comparison results of therespective states. It should be noted that the path metric isinitialized by an initial setting circuit 43 when this circuit isinitiated.

[0059] Next, in FIG. 16, there is shown one embodiment to a magneticrecording/reproducing apparatus with employment of the data demodulatingcircuit of the present invention. An external apparatus such as apersonal computer transmits/receives data via a controller 102 providedin the magnetic recording/reproducing apparatus.

[0060] First, a description will now be made of such a case that datatransmitted from the external apparatus is recorded. Upon receipt of adata recording instruction, the controller 102 issues an instruction toa servo control circuit 103 such that a recording/reproducing head 4 ismoved to a position to be recorded (namely, track). After the transportof the recording/reproducing head is accomplished, recording data issupplied via a recording data processing circuit 104, an R/W amplifier5, and a recording/reproducing head 4 to a recording medium 3 so as tobe recorded on this recording medium 3.

[0061] The recording data processing circuit 104 is arranged by anencoder 23-1, a synthesizer 112, a precorder 9, and a record correctingcircuit 114. The encoder 23-1 executes a coding process operation of therecording data in accordance with a coding rule, for example, an 8/9 GCR(0, 4/4) code conversion. An encoded data stream is sent out in responseto the recording bit period of the synthesizer 112. Since the precorder9 gives a predetermined constraint condition to the data stream, thedata stream is again code-converted. The record correcting circuit 114eliminates the nonlinear characteristic of the recording processoperation specific to the magnetic recording operation. The recordingprocess operation is carried out by executing the above-describedoperations.

[0062] Next, a data reproducing operation will now be described. uponreceipt of a data reproducing instruction, the controller 102 issues aninstruction to the servo control circuit 103 such that therecording/reproducing head 4 is moved to a position on which data hasbeen recorded (namely, track). After the movement of therecording/reproducing head 4 has been completed, a signal recorded onthe recording medium 3 is inputted via the recording/reproducing head 4and the R/W amplifier 5 to the data demodulating circuit 1. Thedemodulation data demodulated by the data modulating circuit 1 isoutputted to the controller 102. After the controller 102 confirmscorrectness of the demodulation data, the controller 102 transfer thedemodulation data to the external apparatus.

[0063] The data demodulating system is arranged by the AGC circuit formaking the amplitude of the head reproduction waveform constant/theband-eliminating filter (LPF) 15 for eliminating the noise outside thesignal band; the ADC 16 for sampling the reproduction signal; theequalizer 10 for eliminating the interference among the codes of thereproduction waveform; the PLL 20 for determining the sampling timing ofthe ADC 16; the data demodulating circuit 1 functioning as a majorcircuit of the present invention and the decoder 23-2 for performing thedecoding process (8/9 GCR decoder) of the demodulation data. Themicrocomputer 101 executes the process operations of the overallapparatus such as the controller 102 and the data demodulating circuit1.

[0064] In this case, the microcomputer 101 executes the followingprocess operations. That is, a detection of a detection result of anirregular code detecting circuit 128, and a setting operation is made ofa register 130 for applying information to a multiplexer 129 forswitching a PRML processing unit 22 and an MEEPRML processing unit 19.Furthermore, the data demodulating system may be alternatively arrangedby adoptively switching these circuits in response to the recordingdensity by employing another MEEPRML circuit having the coefficientlisted in the table 1. This alternative arrangement may be realized bysetting a desirable coefficient of the discrete time filter 18 to theregister 131 by way of the microcomputer 101.

[0065] Moreover, as previously described, in accordance with the presentinvention, since either a 1-bit length or a 3-bit length predominantlyconstitutes the lengths of errors produced in the output data from themaximum likelihood decoder, the error correction suitable for this errorlength is carried out. Thereafter, the decoding process operation suchas 8/9 GCR is performed by the decoder 23. This error correction and thedecoding process operation are preferable so as to prevent the errorcodes from being enlarged. To this end, when the LSI of the datademodulating circuit 1 according to the present invention isconstructed, there is an advantage that such a wiring line 132 is madeat the LSI output terminal in such a manner that the output of themultiplexer 129 is separated into two lines, and the output of themaximum likelihood decoder before the decoding operation is directlyoutputted.

[0066] As previously explained, in accordance with the embodiment, sincethe regenerative isolated magnetized inverse waveform of the magneticrecording apparatus is changed into the asymmetrical waveform, the errorpropagation occurred in a specific pattern which causes the majorproblem in EEPRML and EEEPRML may be suppressed. As compared with theEEPRML system, the MEEPRML system may achieve an improvement of the SINratio higher than, or equal to approximately 1.5 dB in such a case thatthe ratio of a half bandwidth of the reversal of regenerativeisolated-magnetization of the magnetic recording apparatus to a halfbandwidth of the recording signal is on the order of 2.5, namely withinthe practical range of the magnetic recording apparatus. In accordancewith the present invention, furthermore, the length of the code errormay be improved. That is, either a single bit error or a 3-bit 5 errormay mainly occur, as compared with the long/continuous bit errorsoccurred in the conventional EEPRML system and also the conventionalEEEPRML system.

What is claimed is:
 1. A data demodulating method comprising the stepsof: demodulating predetermined input data based upon a responsecharacteristic of the partial response class 4; discrete-filtering saiddemodulated input data to thereby produce filtering data; andmaximum-likelihood-decoding said filtering data to thereby producingasymmetrical response data.
 2. A data demodulating method as claimed inclaim 1 wherein: said discrete-filtering step is a phase filtering stepto produce a minimum phase transition waveform.
 3. A data demodulatingmethod as claimed in claim 1 wherein: said discrete-filtering stepproduces an asymmetrical response of an integer coefficient.
 4. A datademodulating method as claimed in claim 3 wherein: said integercoefficient causes a response of a data stream to become such a responsehaving an integer coefficient that a value obtained by dividing adistance between signals by a temporal relative product of noise powerand noise becomes maximum.
 5. A data demodulating method as claimed inclaim 1 wherein: the demodulation data is applied to the EEPRML system,and a single pulse response of a preselected integer coefficient is aratio of normalized signal values as to any one set of (5, 4, −3, −4,−2), (2, 2, −1, −2, 1), and (3, 2, −2, −2, −1).
 6. A data demodulatingmethod as claimed in claim 1 wherein: the demodulation data is appliedto the EEEPRML system, and a single pulse response of a preselectedinteger coefficient is a ratio of normalized signal values as to any oneset of (2, 5, 1, −3, −2, −2), and (2, 4, 0, −3, −2).
 7. A datademodulating method as claimed in claim 1, further comprising: a stepfor combining with any one of a 1-bit consecutive error and a 3-bitcontinuous error with respect to a code-error correction code.
 8. A datademodulating apparatus comprising: processing means for processing apredetermined input data stream based upon the partial response class 4;a discrate time filter for discrete-filtering the output of saidprocessing means; a maximum likelihood decoder for maximumlikelihood-decoding the output of said discrate time filter to therebyexecute data discrimination; and demodulating means for demodulating theoutput of said maximum likelihood decoder.
 9. A data demodulatingapparatus as claimed in claim 8 wherein: both said discrate time filterand said maximum likelihood decoder are arranged by a maximum likelihooddecoder of the partial response class
 4. 10. A data demodulatingapparatus as claimed in claim 8 wherein: said discrate time filter isconstituted by an analog filter for changing the output of saidprocessing means into a response to a minimum phase transitioncondition.
 11. A data demodulating apparatus as claimed in claim 8wherein: said discrate time filter is constituted by a digital filterfor converting the output of said processing means into an asymmetricalresponse of an integer coefficient.
 12. A data demodulating apparatus asclaimed in claim 13 wherein: said digital filter selects an integervalue in such a manner that a response to a data stream becomes aresponse of an integer series such that a value obtained by dividing adistance between signals by a temporal relative product between noisepower and noise becomes maximum.
 13. A data demodulating apparatus asclaimed in claim 12 wherein: the demodulation data is applied to theEEPRML system, and a single pulse response of a preselected integercoefficient is a ratio of normalized signal values as to any one set of(5, 4, −3, −4, −2), (2, 2, −1, −2, −1), and (3, 2, −2, −2, −1).
 14. Adata demodulating apparatus as claimed in claim 12 wherein: thedemodulation data is applied to the EEPRML system, and a single pulseresponse of a preselected integer coefficient is a ratio of normalizedsignal values as to any one set of (2, 5, 1, −3, −3, −2), and (2, 4, 0,3, −2).
 15. In a magnetic recording/reproducing apparatus forrecording/reproducing an input data stream, the data reproducingapparatus as described in claim 8 is provided in a reproducing system.16. A magnetic recording/reproducing apparatus as claimed in claim 15,further comprising: an external register connected to said discrate timefilter, for setting a coefficient of said discrate time filter.